1. Field of the Invention
The present invention is directed generally to a method for identifying the presence or absence of a known frequency in an input signal composed of a plurality of frequencies, and in particular to a method supplying an input signal to one signal input of each of a plurality of mixers and supplying a further input signal derived from a stable reference frequency to the remaining signal inputs of the mixers wherein the outputs of the mixers are filtered and amplitude evaluated.
2. Description of the Related Art
An example of the need for identifying the presence or absence of defined frequencies having a known value in an input signal composed of a plurality of frequencies is the decoding and recognition of program type information in a two-carrier television audio system. A two-carrier television audio system is described in some detail in the publication entitled "Transmission of Two or More Sound Programmes or Information Channels in Television,"Report 795-1, CCIR Documents, pages 205-213 (1978-1982). In accordance with that disclosure, the program type recognition in the two-carrier television audio system occurs with the assistance of an amplitude-modulated pilot carrier tone. The frequency of the pilot carrier tone is equal to 3.5 times the video line frequency f.sub.h . An identifying tone is modulated onto the pilot carrier tone for the stereo and duel two channels, or tone, operating modes. The frequency spectrums for the possible operating modes are shown in FIGS. 1A, 1B, 1C and 1D.
FIG. 1A shows a frequency spectrum for a stereo operating mode in which a pilot carrier tone having a frequency f.sub.p and that is equal to 3.5.times.15,625 Hz as an identifying tone at a frequency of 15,625/133 Hz modulated onto it by amplitude modulation. The video line frequency f.sub.h thus has a value of 15,625 Hz. In FIG. 1A the side-bands for the stereo signals are identified as f.sub..times.s and f.sub.-s.
In FIG. 1B is shown a frequency spectrum for a two tone, or two channel, operating mode, such as SAP mode, wherein an identifying tone having a frequency of 15,625/57 Hz is modulated onto the pilot carrier tone having a frequency of f.sub.p. In this case, the side-bands are referenced f.sub..times.z and f.sub.-z.
FIG. 1C shows a mono operating mode in a two-carrier television audio system wherein only the pilot carrier-tone having a frequency of f.sub.p is present on which no identifying tone is modulated.
Finally, FIG. 1D shows a standard mono operating mode in the previously standard television transmission mode wherein no pilot carrier tone is present.
The degree of modulation in the operating modes shown in FIGS. 1A and 1B amounts to 50%.
A transmission of the amplitude of the modulation frequencies set forth above ensues with a television transmission signal such that the amplitude modulated pilot carrier tone is modulated onto a carrier in the MHz domain with frequency modulation. In the television receiver, the frequency demodulation is first carried out and this is followed by an amplitude demodulation to acquire the identifying tones. The interpretation of the mono operating mode, the two channel operating mode, and the stereo operating mode can occur via the upper or lower sideband. The interpretation is thereby equivalent.
Up to now, the decoding of the program recognition in two-carrier television audio systems is usually carried out by sync or envelope rectification of the modulated pilot carrier tone with a subsequent interpretation of the demodulated identified tones.
In the transmission of television information, any unwanted frequencies that are identical to the frequency of the pilot carrier tone can arise from the video part of the television information as the result of cross-modulation effects. These unwanted frequencies can be so strong that corresponding pronounced fluctuations of the amplitude of the pilot carrier tone and, thus, undesired, unwanted frequencies appear after the amplitude demodulation.
In the publication by Kuemmel, "Zwei-Kama;-Funkschau", No. 2 76-79, (January 1982), is disclosed an automatically switchable stereo-dual tone decoder provided for a two-carrier television audio system, the switching information needed for the identification of the operating mode is acquired through the use of two identical phase locked loop circuits that are tuned to the respective identifier frequency by a potentiometer.
In European published patent application EP-OS No. 0 146 749 is disclosed a circuit arrangement operating according to a different principle.
FIG. 2 shows a block circuit diagram for a circuit arrangement of the prior art which is taken from the European published application No. 0 146 749;
FIG. 3 shows a block circuit diagram of an embodiment of a frequency generator which is provided in the circuit arrangement of FIG. 2; and
FIGS. 4A and 4B are diagrams for explaining the interpretation of dc voltage parts corresponding to a frequency to be recognized.
In the circuit arrangement of FIG. 2, an input signal is supplied to an input 20. The presence or absence of a frequency of a known value in the input signal is to be determined. For the sake of simplicity, let the input signal be represented by: EQU A sin (Omega t+fi),
Wherein A denotes the amplitude of the signal, Omega denotes its frequency, t denotes the time, and fi denotes the phase of the input signal.
Two switch signals having a frequency identical to the frequency to be identified, also referred to as the frequency of interest, and having a phase difference of 90.degree. are generated by a frequency generator 21 from a stable reference frequency which is supplied to an input 22. In the case of a two-carrier television audio system, it is expedient to use the video line frequency f.sub.h , which is available in the video signal, as the stable reference frequency.
The input signal at the input 20 is supplied to one input of each of two liner mixers 25 and 26. One of the two switch signals generated by the frequency generator is supplied to a further input of the mixer 25 via a line 23 while the other of the two switch signals generated by the frequency generator 21 is supplied to the other mixer 26 via a line 24. A multiplicative mixing of the input signal with the respective switch signals ensues in the mixers 25 and 26. When the frequencies of the input signal and of the two switch signals are identical, then the outputs of the mixers 25 and 26 still only supply a dc voltage component X or Y, respectively, that is proportional to the amplitude A and the phase angle fi of the input signal. Therefore, the following approximations are valid. EQU X=A sin fi EQU Y=A cos fi
As may be seen, the input signal is evaluated in two channels with a phase difference of 90.degree. between the switch signals supplied to the mixers 25 and 26, since the phase fi of the input signal may be arbitrary, i.e. may be either 0.degree. or 90.degree. , or a multiple thereof by a factor of 180.degree. . Thus, the dc voltage part according to the two equations recited above may disappear given evaluation in one channel.
The two mixers 25 and 26 are each followed by a filter in the form of low-pass filters 27 and 28 that serve the purpose of filtering out ac components from the output signals of the respective mixers 25 and 26. Each of the low-pass filters 27 and 28 has a cut-off frequency of a few Hz, and preferably at most two Hz. With respect to the input signal at input 20, the low-pass filters 27 and 28 act like a band pass filter having a center frequency equal to the switching frequency of the two switch signals appearing on lines 23 and 24 and having a band width equal to twice the cut-off frequency of the low-pass filters 27 and 28. Only the dc voltage components X and Y therefore appear at the outputs of the low-pass filters 27 and 28 given identity of the frequency of the signal at the input 20 and of the switch signals on the lines 23 and 24.
The output signals from the low-pass filters 27 and 28 are each supplied to a corresponding amplitude evaluator 29 and 30 with which the amplitude of the dc voltage component X or Y is evaluated. An interpretation of the amplitude suffices for identifying the presence or non-presence, i.e. absence, of a frequency of a known-value in the input signal at the input 20 since the phase fi of the input signal is arbitrary.
The amplitude evaluation can ensue in various ways. For example, the amplitude evaluators 29 and 30 can be formed as threshold switches each having a threshold input 33 and 34 that identify whether the absolute values of the dc voltage components X and Y are higher than a prescribed threshold value S which is fed to the threshold inputs 33 and 34. This type of amplitude evaluation is shown in FIG. 4A. The dc voltage component is thereby in turn established by an amplitude and by a resultant angle so that the dc voltage component is formed by a phaser, or indicator, 50 that lies on a circle 51 depending upon the arbitrary angle .PHI. of the signal. The amplitude evaluators 29 and 30 of FIG. 2 which are formed as threshold switches respond to a threshold S which results in a square 52 having an edge length corresponding to twice the threshold of the threshold switches 29 and 30 for evaluation of the dc voltage component having an amplitude A and a resultant angle .PHI.. When the indicator 50 for the dc voltage component corresponding to the amplitude A lies outside the square 52 defined by the threshold S, then the dc voltage component is output at the outputs 31 and 32 of the amplitude evaluators 29 and 30 of FIG. 2, which establishes that the presence of the frequency to be identified has been measured in the input signal at the input 20.
A further possibility for evaluating the dc voltage components X and Y is by way of calculating the square root of the sum of the squares with conventional electronic elements and identifying whether the value of this root is greater than a prescribed threshold value S. This situation is shown in FIG. 4B, in which similar elements are given identical reference numbers. An evaluator circle 54 is present instead of the evaluator square 52 of FIG. 4A because the square root of the sum of the squares having the values X and Y traces out a circle. As above, the presence of a dc component lying outside the evaluator circle 54 results in the dc voltage component appearing at the outputs of the evaluator circuits. Of course, in this embodiment, the threshold inputs pf the evaluator circuits have the parameters for the evaluator circle of radius S in place of the angle constant values of the example in FIG. 4A.
FIG. 3 shows an embodiment of a frequency generator 21 in the form of a phase locked loop (PLL) circuit for generating a switch signals for the mixers 25 and 26 in FIG. 2. A voltage-controlled oscillator (VCO) 40 oscillates at the frequency of the switch signal for the mixers 25 and 26. The output signal of the oscillator 40 is output on a line 41. A stable reference frequency which is supplied into the input 22 of FIG. 2 is supplied to the corresponding input 42 in the circuit of FIG. 3. For frequency matching, a frequency divider 43 follows the input 42 and a frequency divider 44 follows the voltage-controlled oscillator 40. The output signals from the frequency dividers 43 and 44 are compared to one another in a phase comparison stage 45. The output signal from the phase comparison stage 45 is supplied to the voltage controlled oscillator 40 through a low-pass filter 46, which results in frequency control of the oscillator 40.
For generating the two switch signals appearing on lines 23 and 24 for the mixers 25 and 26, the switch signals are phase-shifted by 90.degree. to one another. The output line 41 of the phased locked loop circuit of FIG. 3 is divided at a phase-shifter 47 which shifts the output signal by 90.degree. in one of the two divided output lines 23 and 24. Such a phase-shifter circuit 47 is inherently conventional, and since it is not part of the phase-locked loop circuit, it is shown in FIG. 3 in dotted outline.
In the specific case of two-carrier television audio system in the PAL system, the frequency divider 43 divides the horizontal sweep frequency f.sub.h of 15,625 Hz by 133 in the case of the stereo operating mode and by 57 in the case of a two channel, i.e. SAP, operating mode. For stereo operation, the frequency of the upper sideband f.sub.+s to be detected thus derives as EQU f.sub.+s =7/2 f.sub.h +f.sub.h /133 or EQU f+s=933/266 f.sub.h.
Given the horizontal scan frequency f.sub.h /133, the identifier tone in the stereo operating mode lies at 117 Hz. Phase differences between the output signal of the frequency divider 43 and the output signal of the frequency divider 44 lead to control spikes at the output of the phase comparison stage 45. When the cut-off frequency of the low-pass filter 46 used as the control filter is not low enough or when its attenuation at 117 Hz does not lie on the order of magnitude of 30-40 dB, a frequency modulation at the oscillator 40 is obtained on the basis of these control spikes with sidebands that correspond to a present stereo signal, even if no pilot tone is present. The phase locked loop circuit of FIG. 3 becomes extremely sluggish due to a low-off frequency for the low-pass filter 46 corresponding to the requirements and due to a correspondingly small loop band width. Appropriate dimensioning of the phase locked loop circuit is extremely critical and the leakage currents in its control loop dare not lie in the nano amp range.
Due to a correspondingly slow frequency editing circuit, however, the overall circuit arrangement as disclosed in European published application No. 0 146 749 becomes extremely sluggish. This loop band width allows an admissable deviation in the horizontal frequency in the order of magnitude of only about 1Hz. For example, in video tape recordings, a greater horizontal frequency deviation can arise as a consequence of synchronization fluctuations in the recording or the playback means, so that an undisturbed detection of the spectral line of a characteristic frequency of the audio modes is not possible with such known circuit arrangements.